The use of LEDs in illumination systems are well known as they offer significant advantages over traditional light sources such as higher efficacy, increased reliability due to their solid-state nature and increased longevity amongst many other advantages known to those familiar in the area of LEDs and OLEDs.
(O)LEDs are used in a wide variety of configurations for general and specific illumination applications including, but not limited to task lighting, accent lighting, emergency lighting, hospitality lighting, restaurant lighting, hospital lighting, office lighting, retail lighting, automotive lighting, street lighting, amenity lighting, effect lighting, marine lighting, display case lighting, TV, film and projection lighting, entertainment lighting, animal and food production lighting, medical lighting, outdoor lighting and backlighting of displays, corridor lighting, and the like.
LEDs and OLEDs are current-controlled devices where the intensity of light emitted from the device is related to the amount of current driven through the device. It is therefore highly advantageous to carefully and reliably control the amount of current flowing through the LED or OLED device(s) in order to achieve the desired illumination effect from an illumination system and to maximise the life of a device by ensuring the maximum current or power specifications are not exceeded.
(O)LED power supply systems have been developed based on a variety of circuit design topologies which provide the ability to vary the actual or time-averaged forward current through the light emitting device load over an acceptable range in order to provide dimming capabilities. (O)LED illumination systems have been devised which, through the use of multiple light emitting devices having discrete wavelengths/colours, can produce a variety of colours and intensities. Systems incorporating Red, Green, Blue, Amber and White light emitters can create near infinite colour variations by varying the intensity, current or power of each of the coloured light emitter(s) individually or together in combination.
As (O)LED illumination systems have become accepted by the general lighting industry various methods have been devised to control the current driven through the light emitting devices under control to achieve the desired dimming and colour mixing.
One common method for modulating the current through (O)LED devices is Pulse Width Modulation (PWM) such as that highlighted in patents GB2398682A, U.S. Pat. Nos. 6,305,818B1, 6,016,038, US2004/0155844A1, US2005/0218838A1, WO02/48994A1, GB2346004A and U.S. Pat. No. 5,008,595, all of which are incorporated herein by reference as if set forth in full. A PWM driver provide a pulsing of the current(s) through one or more (O)LEDs by changing from full current “ON” state to a zero current “OFF” state and is often know as digital dimming. The duty cycle of a PWM driver is defined by the ratio of the ON time to the total cycle time in a fixed cycle frequency. Dimming or power control of the (O)LEDs may be achieved by varying the duty cycle of current through the light emitting device(s) from 100% to 0% as the human eye integrates the ON/OFF pulses into a time averaged luminous intensity.
Although PWM schemes are common place due to the simplicity of digital switching between two binary states of ON and OFF they have significant disadvantages including:                1. Uneven power supply loading: By switching the light emitting device loads on and off at the same time causes the power supply to provide zero output or full output on a continuously switched basis which reduces the overall efficiency of the power supply unit (PSU)        2. The Electromagnetic interference (EMI) of a PWM system is far more complicated than with other control methods as the changes in duty cycle results in a wide frequency range of noise being emitted.        3. The amplitude of the EMI is increased as the forward current is increased as the pulsing currents through the electrical leads connecting the driver electronics to the illumination system fixtures may act as transmitters of radio frequency energy that could cause interference with other equipment as the interference is very high at the transition of the current switching.        4. The pulsing of PWM systems for a variety of applications are not acceptable especially where high refresh rates are required. For example, in TV and Film environments the camera's are usually digital and a mismatch between the camera capture frequency and the PWM fixed frequency results in poor quality imaging in both intensity and colour.        5. The dimming dynamic range of PWM systems are usually very poor as the only way to dim the lighting loads is by reducing the duty cycle and at low duty cycles there is a very small percentage of time when the light emitting device is actually illuminated. Some PWM systems offer high resolution 16-bit dimming (1:65535) ratios however as the dimming range is extended the maximum fixed frequency has to reduce which can results in visual flicker or a beating effect.        6. PWM systems have a fixed maximum forward current and additional circuitry is required in order to change this adding extra cost and complexity to the design.        7. PWM based systems are much less efficient at driving (O)LEDs when being dimmed in some circumstances up to 150% less efficient compared to DC or constant current reduction dimming as discussed by Gu, Y et al, in the SPIE 2006 paper entitled “Spectral and Luminous Efficacy Change of High-power LEDs under Different Dimming Methods”. SPIE 6337, 63370J.        8. Additional thermal and electrical stresses are placed on the lighting load as power is rapidly cycled ON and OFF causing thermal cycling that adds stress to the LED die or bond wires attached to the LED die.        9. PWM systems make it more difficult to control complete illumination systems as it only has one variable (Duty Cycle) in which to change the control system output driver stage(s). For example, if an external thermal input is required into the control system additional duty cycle resolution will be required in order to control the output driver stage(s) according to the dimming level as a function of the illumination fixture temperature.        
A second method known as Pulse Amplitude Modulation (PAM) and highlighted in patents GB 2369730 and GB 2408315A have been developed to overcome many of the shortfalls of PWM driver systems and is a hybrid dimming technique between pulsing and analogue topologies. Such a system offers improved dimming, efficacy and control performance by enabling two variable parameters to be effectively used to control current and power through the light emitting device(s) (pulse width and amplitude of pulse).
A third method often known as Direct Current (DC) dimming or Constant Current Reduction (CCR) has been used to power light emitting devices however the difficulty with such a technique is to obtain high efficiency from the driver output stages as they are usually implemented with linear power supplies or the dynamic dimming range is not high (1:100 or 1:200). In addition, linear power supplies suffer from efficiency drops as the driver system input voltage deviates from that of the voltage required to power the emitting device(s). Therefore, the bigger the voltage difference the more power is wasted as heat across the linear power supply components which, in turn, can cause early failure of components and reduces overall driver lifetime and efficiency.
Further current driving methods are highlighted in a variety of industry literature to include Pulse Frequency Modulation (PFM) that fixes the pulse width and varies the frequency and variants such as Pulse Density Modulation (PDM), Delta Sigma Modulation (DSM) and Stochastic Signal Density Modulation (SSDM) all of which enable dimming to one degree or another.
Irrespective of the output drive current technique a key feature of driving solid-state light sources is the ability to dim the light emitting device(s) to very low intensity levels using precise current control within an illumination system containing a single wavelength/colour, multiple discrete wavelengths/colours, broadband wavelength (white) or a combination of broadband and single wavelength(s). A significant issue of the latest generation of (O)LED devices is that their efficiency to produce light from electrical current is so high that even at very low levels of current ie; 1-10 mA the light being emitted is considerably brighter than desired in many applications due to the sensitivity of the human eye. Many of the latest O(LED) devices can be operated with forward currents up to 36 A (eg; Luminous Devices Inc LEDs) and so a wide dynamic dimming range is required in order to provide smooth and accurate dimming at all light levels.
The majority of solid state lighting power control systems are designed to offer fixed current or output voltage parameters in order to maximise total system efficiency. For example, the majority of LED drivers today are categorised by the maximum forward current eg; 350 mA, 700 mA or 1000 mA operating over a defined output voltage range eg; 30-48V. These topologies are inflexible such that if different (O)LED light emitting device(s) are used with varying characteristics the performance of the power control system will at best be altered significantly but at worst not function correctly at all. A further aspect of the present invention relates to improvements in methods and apparatus to optimise the efficiency and flexibility of the power control circuit to the parameters of the light source(s) connected enabling embodiments to optimise performance, lifetime and widen the scope of use.
Furthermore where the illumination system fixture requires independently controlled light source channels such as in Red, Green, Blue and Amber the actual power used across all of the channels is significantly less than the maximum power for the combined channels and this results in the driver system operating at less than optimal performance. A further aspect of the present invention relates to dynamically sharing of power across multiple output channels controlling different light source(s) to optimise the output power of the power control circuit.
Typical power control topologies employed to deliver power to light source(s) include:                Linear Regulators: The linear regulator is the basic building block of nearly every power supply used in electronics. The IC linear regulator is very easy to implement.        Switch Mode Power Supplies (SMPS): are an electronic power supply incorporating a switching regulator in order to be highly efficient in the conversion of electrical power.        
There are three basic types of linear regulator designs used in electronics:                1. Standard (NPN Darlington) Regulator        2. Low Dropout or LDO Regulator        3. Quasi LDO Regulator        
SMPSs can be classified into four types according to the input and output waveforms:                AC in, DC out: rectifier, off-line converter input stage        DC in, DC out: voltage converter, or current converter, or DC to DC converter        AC in, AC out: frequency changer, transformer, phase converter        DC in, AC out: inverter        
Furthermore, switched-mode power supplies can be classified according to the switching circuit topology. The most important distinction is between isolated converters and non-isolated ones. Typical non-isolated circuit topologies include Buck, Boost, Buck-Boost, Split-Pi, Cuk, SEPIC, Zeta and Charge pump types. Typical isolated circuit topologies include Flyback, Ringing Choke Convertor, Half-forward, Forward, Resonant Forward, Push-Pull, Half-Bridge, Full-Bridge, Resonant Zero Voltage Switched and isolated Cuk.
The two types of regulators have their different advantages:                Linear regulators are best when low output noise (and low RFI radiated noise) is required        Linear regulators are best when a fast response to input and output disturbances is required.        At low levels of power, linear regulators are cheaper and occupy less printed circuit board space.        Switching regulators are best when power efficiency is critical except linear regulators are more efficient in a small number of cases (for example if the complexity of the switching circuit and the junction capacitance charging current means a high quiescent current in the switching regulator).        Switching regulators are required when the only power supply is a DC voltage, and a higher output voltage is required.        At high levels of power (above a few watts), switching regulators are cheaper (for example, the cost of removing heat generated is less).        
The prior art circuit in FIG. 1a gives an example of a typical buck power stage controller schematic usually employed to power solid-state light emitting device(s). During normal operation of the buck power stage, Q1 is repeatedly switched on and off with the on and off times governed by the control circuit. This switching action causes a train of pulses at the junction of Q1, CR1, and L which is filtered by the L/C output filter to produce a DC output voltage, VO.
A power stage can operate in continuous or discontinuous inductor current mode. Continuous inductor current mode is characterized by current flowing continuously in the inductor during the entire switching cycle in steady state operation. Discontinuous inductor current mode is characterized by the inductor current being zero for a portion of the switching cycle. It starts at zero, reaches a peak value, and returns to zero during each switching cycle. It is very desirable for a power stage to stay in only one mode over its expected operating conditions, because the power stage frequency response changes significantly between the two modes of operation.
The voltage conversion relationship for the continuous conduction mode buck power stage shows how the output voltage depends on duty cycle and input voltage or, conversely, how the duty cycle can be calculated based on input voltage and output voltage.
In continuous conduction mode, the Buck power stage assumes two states per switching cycle. The ON state is when Q1 is ON and CR1 is OFF. The OFF state is when Q1 is OFF and CR1 is ON. A simple linear circuit can represent each of the two states where the switches in the circuit are replaced by their equivalent circuits during each state. The circuit diagram for each of the two states is shown in FIG. 1b and 1c. 
The duration of the ON state is D×TS=TON where D is the duty cycle, set by the control circuit, expressed as a ratio of the switch ON time to the time of one complete switching cycle, Ts. The duration of the OFF state is called TOFF. Since there are only two states per switching cycle for continuous mode, TOFF is equal to (1−D)×TS. The quantity (1−D) is sometimes called D′. These times are shown along with the waveforms in FIG. 2.
Referring to FIG. 1b, during the ON state, Q1 presents a low resistance, RDS(on), from its drain to source and has a small voltage drop of VDS=IL×RDS(on). There is also a small voltage drop across the dc resistance of the inductor equal to IL×RL. Thus, the input voltage, VI, minus losses, (VDS+IL×RL), is applied to the left-hand side of inductor, L. CR1 is OFF during this time because it is reverse biased. The voltage applied to the right hand side of L is simply the output voltage, VO. The inductor current, IL, flows from the input source, VI, through Q1 and to the output capacitor and load resistor combination. During the ON state, the voltage applied across the inductor is constant and equal to VI−VDS−IL×RL−Vo. Adopting the polarity convention for the current IL shown in FIG. 1, the inductor current increases as a result of the applied voltage. Also, since the applied voltage is essentially constant, the inductor current increases linearly. This increase in inductor current during TON is illustrated in FIG. 2. The amount that the inductor current increases can be calculated by using a version of the familiar relationship:
      V    L    =                    L        ×                              ⅆ                          i              L                                            ⅆ            t                              ⇒              Δ        ⁢                                  ⁢                  I          L                      =                            V          L                L            ×      Δ      ⁢                          ⁢      T      The inductor current increase during the ON state is given by:
      Δ    ⁢                  ⁢                  I        L            ⁡              (        +        )              =                              (                                    V              I                        -                          V              DS                        -                                          I                L                            ×                              R                L                                              )                -                  V          O                    L        ×          T      ON      The quantity, ΔIL, is referred to as the inductor ripple current.
Referring to FIG. 1c, when Q1 is OFF, it presents a high impedance from its drain to source. Therefore, since the current flowing in the inductor L cannot change instantaneously, the current shifts from Q1 to CR1. Due to the decreasing inductor current, the voltage across the inductor reverses polarity until rectifier CR1 becomes forward biased and turns ON. The voltage on the left-hand side of L becomes −(Vd+IL×RL) where the quantity, Vd, is the forward voltage drop of CR1. The voltage applied to the right hand side of L is still the output voltage, VO. The inductor current, IL, now flows from ground through CR1 and to the output capacitor and load resistor combination. During the OFF state, the magnitude of the voltage applied across the inductor is constant and equal to (VO+Vd+IL×RL). Maintaining the same polarity convention, this applied voltage is negative (or opposite in polarity from the applied voltage during the ON time). Hence, the inductor current decreases during the OFF time. Also, since the applied voltage is essentially constant, the inductor current decreases linearly. This decrease in inductor current during TOFF is illustrated in FIG. 2.
The inductor current decrease during the OFF state is given by:
      Δ    ⁢                  ⁢                  I        L            ⁡              (        -        )              =                              V          O                +                  (                                    V              d                        +                                          I                L                            ×                              R                L                                              )                    L        ×          T      OFF      This quantity, ΔIL(−), is also referred to as the inductor ripple current. In steady state conditions, the current increase, ΔIL(+), during the ON time and the current decrease during the OFF time, ΔIL(−), must be equal. Otherwise, the inductor current would have a net increase or decrease from cycle to cycle which would not be a steady state condition. Therefore, these two equations can be equated and solved for VO to obtain the continuous conduction mode buck voltage conversion relationship.Solving for VO:VO=(VI−VDS)×D−Vd×(1−D)−IL×RL In the above equations for ΔIL(+) and ΔIL(−), the dc output voltage was implicitly assumed to be constant with no AC ripple voltage during the ON time and the OFF time.
The above voltage conversion relationship for Vo illustrates the fact that VO can be adjusted by adjusting the duty cycle, D, and is always less than the input because D is a number between 0 and 1. A common simplification is to assume VDS, Vd, and RL are small enough to ignore. Setting VDS, Vd, and RL to zero, the above equation simplifies considerably to:VO=VI*D. 
To relate the inductor current to the output current, referring to FIGS. 1 and 2, note that the inductor delivers current to the output capacitor and load resistor combination during the whole switching cycle. The inductor current averaged over the switching cycle is equal to the output current. This is true because the average current in the output capacitor must be zero. In equation form, we have:IL(Avg)=IO 
Although buck regulators and other switching topologies can be operated in discontinuous conduction modes the preferred operation for the majority of drivers within the marketplace is the continuous conduction mode. Therefore, switching topologies are not suitable for precise or repeatable current control at low forward currents and the lowest current limit of a system is determined by the switching speed (or minimum pulse width) achieved by the MOSFET switching device. In order to improve the dynamic current range of switching devices one method set forth in US patent 2009/0302779A1 combines an analogue feedback signal into a switching buck stage from 100% current dimming down to approximately 10% current with a PWM varying duty cycle to dim from 10% to 0%. Although this technique achieves a wider dynamic dimming range compared to a standard switching technique there are still pulses in the current through the light emitting device(s) load and maintains the majority of disadvantages attributed to the PWM dimming technique at lower currents. Also the circuit topology described is configured such that it is not possible to operate said power system in a common anode topology for more than one light emitting device output. In addition, the circuit topology described is not able to bond one or more output channels together to increase total (O)LED output current. A further disadvantage of US patent 2009/0302779A1 is that the design does not contain fault tolerant error protection so if the anode terminal is shorted to ground or to the cathode of another output connector the switching IC or MOSFET would fail rendering the power controller inoperable.
Further improvements over US patent 2009/0302779A1 have been made whereby the analogue control has been integrated into a single semiconductor IC such as the National Instruments LM3414. The LM3414 utilises two control inputs to modulate LED brightness as shown by the simplified circuit schematic in FIG. 3. An analog current control input, IADJ, is provided so the LM3414 can be adjusted to compensate for (O)LED manufacturing variations and/or temperature correction whilst the PWM output functions by shorting out the (O)LED with a parallel switch allowing high PWM dimming frequencies using the DIM input. Further improvements include common anode capabilities and operation without the need for an external sense resistor to improve efficiency.
The LM3414 operates in a continuous conduction mode and is able to remove the current sense resistor from the design by assuming the average output current of the topology is found half way through the ON or OFF switching period as shown by the Io dashed line in FIG. 2. Although this method is approximately related to the output current it is not an exact representation and if there are any non-linearity's in the inductor current IL or a sample timing issue this assumption is not representative of the average output current. It is also important to note the minimum output current achievable within continuous conduction switching circuits is equal to 50% of the system ripple current or ΔIL/2 otherwise the system will enter into a discontinuous conduction mode. In order to reduce ripple current of the power controller there are several techniques that can be employed individually or combined including:                Increase the switching frequency        Increase the capacitance on the driver output stage        Change the inductance of the inductor        
The switching frequency of the output stage is usually determined during the design phase and is calculated according to the size of components, efficiency of design and EMC/EMI noise requirements. If the switching frequency is too high there are switching losses which reduce the overall efficiency of the stage and EMC/EMI design becomes more complex. Increasing the capacitance on the output stage means bulkier components and if the output voltage is high eg>20V then component costs increase. If the capacitance values exceed a few micro Farad then electrolytic capacitors are required which have shorter lifetimes and are highly temperature dependant compared to ceramic or film based capacitors. Increasing the inductance values of the energy storage component in a switching topology results in higher DC resistances in the inductor which results in lower efficiencies overall and the maximum current of the driver output stage is reduced as the inductance increases for the same size inductors. Therefore, such switching designs are usually limited to being fixed once into production.
In addition, the LM3414 still suffers from the ability to channel bond multiple outputs, dim the output current in an analogue (or non PWM) fashion to low currents (The LM3414 datasheet indicates a minimum analogue dimming current of approximately 100 mA can be achieved) and cannot tolerate miss-wiring of the (O)LED outputs potentially resulting in catastrophic failure of the driver stage in application.
Consequently, linear regulators offer simple design and less EMI issues compared to switching power topologies however their efficiency, size and heat generation limit their scope to lower power applications. A linear regulator provides the desired output voltage by dissipating excess power in ohmic losses (e.g., in a resistor or in the collector-emitter region of a transistor in its active mode). A linear regulator regulates either output voltage or current by dissipating the excess electric power in the form of heat, and hence its maximum power efficiency is voltage-out/voltage-in since the volt difference is wasted.
Switching power supplies are approximately 80%-95% efficient. Higher efficiency usually is an advantage, because heat normally is considered to be wasted energy (at the least) and potentially damaging to nearby electronic components. Like other types of power supplies, an SMPS transfers power from a source like the electrical power grid to a load while converting voltage and current characteristics. An SMPS is usually employed to efficiently provide a regulated output voltage, typically at a level different from the input voltage.
Unlike a linear power supply, the switching transistor or MOSFET of a switch mode power regulator oscillates very quickly (typically between 50 kHz and 4 MHz) between full-on and full-off states, which minimizes wasted energy. Voltage or current regulation is provided by varying the ratio of on to off time. In contrast, a linear power supply must dissipate the excess voltage to regulate the output. This higher efficiency is the chief advantage of a switch-mode power supply.
Switching regulators are used as replacements for the linear regulators when higher efficiency, smaller size or lighter weight are required. They are, however, more complicated, their switching currents can cause electrical noise problems if not carefully suppressed, and simple designs may have a poor power factor.